LT1306 데이터 시트보기 (PDF) - Linear Technology

부품명

상세내역

일치하는 목록

LT1306

Synchronous, Fixed Frequency Step-Up DC/DC Converter

Linear Technology 

LT1306

APPLICATIONS INFORMATION

Since a boost converter produces high output current

ripple, one also needs to consider the maximum ripple

current rating of the output capacitor. Capacitor reliability

will be affected if the ripple current exceeds the maximum

allowable ratings. This maximum rating is usually

specified as the RMS ripple current. In the LT1306 the

RMS output capacitor ripple current is:

IO

VO – VIN

VIN

For 2-cell to 5V applications, 220µF low ESR solid tanta-

lum capacitors (AVX TPS series or Sprague 593D series)

work well. To reduce output voltage ripple due to heavy

load transients or Burst Mode operation, higher capaci-

tance may be used. For through-hole applications, Sanyo

OS-CON capacitors are also good choices.

In a boost regulator, the input capacitor ripple current is

much lower. Maximum ripple current rating and input

voltage ripples are not usually of concern. A 22µF tantalum

capacitor soldered near the input pin is generally an

adequate bypass.

Bootstrap Supply

Diode D1 and capacitor C1 generate a pulsating supply

voltage, VCAP, which is higher than the output. The rectifier

drive circuit runs off this supply. During rectifier on-time,

the rectifier base current drains C1. Q2 base current and

the maximum allowable VCAP ripple voltage determine the

size of C1. A 1µF capacitor is sufficient to keep VCAP ripple

below 0.3V. For a 2-cell input (VIN > 1.8V) over an extended

temperature range, a BAT54 Schottky diode may be used

for D1. The use of a Schottky diode increases the bootstrap

voltage and the operating headroom for the rectifier driver,

X5. Diodes like a 1N4148 or 1N914 work well for 2-cell

inputs over the 0°C to 70°C commercial temperature

range.

The charge drawn from C1 during the rectifier on-time has

to be replenished during the switch on-interval. As duty

cycle decreases, the amplitude of the C1 charging current

can increase dramatically especially when delivering high

power to the load. This charging current flows through the

switch and can cause the current limit comparator to trip

erratically. For boost applications where VIN is a few tenths

of a volt below VO, a 1µF or 2.2µF tantalum capacitor (such

as AVX TAJ series) can be used for C1. The ESR of the

tantalum capacitor limits the charging current. A low value

resistor (2Ω to 5Ω) can also be added in series with C1 for

further limiting the charging current although this tends to

lower the converter efficiency slightly.

Frequency Compensation

Current mode switching regulators have two feedback

loops. The inner current feedback loop controls the

inductor current in response to the outer loop. The outer

or overall feedback loop tightly regulates the output

voltage. The high frequency gain asymptote of the inner

current loop rolls off at – 20dB/decade and crosses the

unity gain axis at a frequency ωc between 1/6 to 2/3 of the

switching frequency. The current loop is stable and is

wideband compared to the overall voltage feedback loop.

The low frequency current loop gain is not high (usually

between unity and 10) but it increases the low frequency

impedance of the inductor as seen by the output filter

capacitor. (In a boost regulator, the inductor is con-

nected to the output during the switch off-time.) Current

mode control introduces an effective series resistance

(>> DCR) to the inductor that damps the LC tank re-

sponse. The complex high-Q poles of the LC filter are now

separated, resulting in a dominant pole determined by

the filter capacitance and the load resistance and a

second high frequency pole.

For a boost regulator the control to output transfer func-

tion can be shown to have a dominant pole at the load

corner frequency

ωP

=



RL

2

1

 (CO )

and a moving right-half plane (RHP) zero with a minimum

value of

ωZ

=

RL(1– DMAX )2

L

10

Share Link: